The present invention relates to a demodulator and, more particularly, to a demodulator for acquiring a carrier frequency for a .pi./4-shift QPSK signal in satellite communications.
Both quadriphase shift keying (QPSK) and .pi./4-shift QPSK are digital signal modulation schemes, in which two carriers, for binary digital signals of two series, which have a phase difference of 90.degree. with respect to each other are subjected to binary phase shift keying to generate two quadrature-modulated waves, and the waves are added to be output, as a QPSK or quadriphase PSK signal, to a transmission path. In both the schemes, a quaternary signal is transmitted. However, in the .pi./4-shift QPSK modulation scheme, one of four phase states on the phase plane of two channel quadrature signals is transmitted at a given symbol period, and one of four phase states set by shifting each of the above four phase states on the phase plane by .pi./4 is transmitted at the next symbol. These transmitting operations are alternately repeated. In this regard, the .pi./4-shift QPSK is different from the QPSK in which such a shifting operation is not performed.
In the .pi./4-shift QPSK modulation scheme, carrier phase states are represented in the manner shown in FIG. 8. As shown in FIG. 8, the phase states are shifted without passing through the origin of a signal space diagram. For this reason, the overall envelope fluctuations of .pi./4-shift QPSK are smaller than those of QPSK in which the envelope goes through zero amplitude when both the in-phase and quadrature channel data change phase simultaneously. Therefore, the spread of spectra can be suppressed even with the use of a nonlinear amplifier.
For the above reason, in using a nonlinear amplifier, such as a class "C" amplifier, which can amplify a signal with a high power efficiency and has a simple structure, the .pi./4-shift QPSK modulation scheme has been conventionally used, which can effectively use a transmission band in accordance with an increase in transmission channel capacity. This .pi./4-shift QPSK has been mainly employed as a modulation scheme in digital portable telephone and cellular mobile telephone systems, and has also been used in mobile satellite communications.
In satellite communications, coherent detection schemes are usually used for greater link margin, unlike in digital portable telephone and cellular mobile telephone systems. In addition, in satellite communications, a carrier frequency offset is caused by the Doppler effect, drifts caused by a local oscillator on the transmission side or on the transponder of a satellite, residual frequency error by automatic frequency control (AFC), and the like. Especially in mobile terminals and earth stations designed to be reduced in size and price, less stable oscillators in frequency need to be used, inviting greater a carrier frequency offset.
Such a carrier frequency offset poses a serious problem in coherent detection. For this reason, when such a large carrier frequency offset is present, various measures are taken on the receiving side to achieve prompt carrier acquisition. For instance, a known demodulator (e.g., disclosed in Japanese Patent Laid-Open No. 3-131149) is used on the receiver side. This demodulator is designed to acquire a carrier frequency by sweeping the oscillation frequency of a voltage-controlled oscillator in a carrier recovery circuit using a phase-locked loop (PLL).
FIG. 9 shows an arrangement of a conventional demodulator. Referring to FIG. 9, the demodulator is basically constituted by a carrier recovery circuit using a phase-locked loop (PLL) constituted by a voltage-controlled oscillator (VCO) 203 for generating a reference carrier, a coherent detector 202 for performing coherent detection of an input signal by using the reference carrier, a phase detector 204 for detecting a phase error between an input signal and the reference carrier, and a loop filter 205 for increasing the signal-to-noise power ratio (S/N) of a recovered carrier. The demodulator also includes a .pi./4 reverse shifter 201 for shifting the phase of an input .pi./4-shift QPSK signal by .pi./4, a unique word (UW) detector 206 for detecting a unique word (synchronizing word or framing) from the data obtained by demodulating the input .pi./4-shift QPSK signal by means of coherent detection, a frame synchronization circuit 207 for establishing/holding synchronization of a reception frame on the basis of the detected unique word, a frequency sweeping circuit 208 for sweeping the output oscillation frequency of the VCO 203, and an adder 209 for adding an output signal from the loop filter 205 and an output signal from the frequency sweeping circuit 208 and applying the resultant signal, as a control voltage, to the VCO 203.
In this conventional demodulator, a received .pi./4-shift QPSK signal is input to the .pi./4 reverse shifter 201, in which the phase of the .pi./4-shift QPSK signal is reversely shifted by .pi./4 for two symbols. The shifted signal is input to the coherent detector 202. The coherent detector 202 then demodulated the QPSK signal from the .pi./4 reverse shifter 201 on the basis of the recovered carrier from the VCO 203, thereby obtaining a demodulated data output of the I- and Q-channel quadrature signals.
This output signal from the coherent detector 202 is supplied to the phase detector 204 to detect the phase error between the output recovered signal from the VCO 203 and the carrier for the QPSK signal from the .pi./4 reverse shifter 201. The phase error is then filtered by the loop filter 205 and applied, as a control voltage, to the VCO 203 via the adder 209. With this operation, the output oscillation frequency of the VCO 203 becomes a recovered carrier conforming/synchronizing to/with the carrier for the QPSK signal coming out of the .pi./4 reverse shifter 201 to the coherent detector 202.
In the above PLL, the sweeping circuit 208 is operated when a carrier is to be acquired after the demodulator is started, and a signal obtained by adding an output signal from the frequency sweeping circuit 208 to an output error signal from the loop filter 205 using the adder 209 is applied, as a control voltage, to the VCO 203.
When the output reference carrier frequency of the VCO 203 is swept and falls within the acquisition range of the PLL, the output reference carrier of the VCO 203 is phase-locked to a carrier for an incoming .pi./4-shift QPSK signal, and the frame synchronization circuit 207 establishes frame synchronization on the basis of a unique word detected by the unique word detector 206 to which demodulated data is input.
The frame synchronization circuit 207 outputs the resultant frame sync signal to the frequency sweeping circuit 208 to stop the frequency sweeping operation of the frequency sweeping circuit 208 and hold its output. With this operation, according to the conventional demodulator, a carrier frequency is acquired, and demodulation is continued.
According to the above conventional demodulator, in the PLL, a carrier phase is synchronized, and frame synchronization is established from demodulated data, thereby completing acquisition of a carrier frequency. For this reason, the acquisition time is dependent on a carrier phase-locking time in the PLL, clock synchronization time, and a frame length, and needs to be shorter than the time defined by these values. Consequently, it takes time to acquire a QPSK signal.
In the conventional demodulator, in order to perform phase lock including a carrier frequency, the PLL must be constituted by a second-order loop. In this case, in a complete integral second-order PLL, a carrier acquisition time T.sub.aq required to synchronize an input signal having a frequency offset .DELTA..omega. is given by ##EQU1## where .lambda. is the damping factor of the loop filter, and .omega..sub.n is the natural angular frequency. It is known that these values and a loop band B.sub.L have the following relationship: ##EQU2## Assume that equation (1) is substituted into equation (1), the damping factor .lambda. is set to shorten the acquisition time T.sub.aq, and .DELTA..omega.&gt;&gt;B.sub.L. In this case, the acquisition time T.sub.aq is given by ##EQU3## That is, the acquisition time T.sub.aq is increased (prolonged) in proportion to the square of the carrier frequency offset .DELTA..omega..
Assume that a low carrier-to-noise power ratio (C/N) is required as an input signal condition. In this case, if the loop band B.sub.L is narrowed to increase the signal-to-noise power ratio (S/N) of a recovered carrier, the carrier acquisition time T.sub.aq is increased (prolonged) in proportion to the cube of the width of the loop band B.sub.L.